Tuesday, November 21, 2017

Matching the 2 and 6 meter cycloid dipoles

The "Cycloid" dipole - a circularly-polarized antenna capable of (more or less) omnidirectional (toward the horizon) radiation was discussed in previous posts:
Figure 1:
The 6 and 2 meter Cycloid dipoles at the WA7X beacon.
The 2 meter cycloid dipole has been in service since 2001 and while the 6 meter
cycloid was (mostly) built at about the same time, it has been in service
only since 2015.
Click on the image for a larger version.
The 2 and 6 meter antennas were installed at the site of the WA7X beacon to impart a circular polarization on the transmitted signals, making them (generally) agnostic to the typical antenna used by someone listening for them - which is to say that it wouldn't matter whether the receive antenna was vertically or horizontally polarized.  The use of circular polarization also reduces the problem where the ionospheric reflection may rotate the signal to the "other" plane and cause fading at the receive location owing to cross-polarization.

While this antenna is described as being "omnidirectional", that is not true in the proper sense of the word.  Its circularity and most of its radiated power is directed toward-ish the horizon (at some elevation angle) in all directions while relatively little energy is radiated upwards - and what is being radiated upwards not likely to be very circular.  As with any antenna, the proximity of the coaxial cable, metallic support and feedline will, no doubt, skew the pattern in some way - and this antenna is no exception - but this is unavoidable.

While the dimensions of the "antenna" part of the Cycloid dipole are spelled out in the linked article(s), above, the matching of these antennas to 50 ohms is not - with only the suggestion that a "1/2 wave matching network" be used.  While this matching network is very simple, it may be unfamiliar to some, so what follows is a paraphrased response to an email on this very question.

Matching the cycloid dipole:

While I carefully noted the dimensions of the dipole when I designed them, we never precisely measured the various dimensions of the matching networks of the 2 and 6 meter Cycloid Dipoles, but they are simply stub-tuned 1/2 wave sections - only the dimensions of the actually "antenna" portion are on the web page.

In retrospect, a full 1/2 wave section was probably an overkill as a 1/4 wave may have sufficed - but that "extra" bit of open-wire balanced transmission line (e.g. the portion of the pipe between the coax tap and the "cycloid" part of the antenna) worked out well to provide physical support, rigidity, and counterbalance, would not cause any significant loss, and it all but guaranteed that we would be able to find a good match.

The details of the matching network and its tuning are thus:
  • We used about 1/2 wave length of copper tubing protruding from the "back" of the antenna, used also to support the main antenna body.  As can be seen from the pictures, it was folded upon itself, zig-zagging to reduce its overall size.  This extra weight can help to counter-balance the weight of the antenna itself.
  • There is a 1/2 wavelength coaxial balun to go from 50 ohms unbalanced to 200 ohms balanced using small 50 ohm coaxial cable..  This type of balun well-described in literature and one of several online calculators may be found here:  http://n-lemma.com/calcs/dipole/balun.htm .  For the 2 meter antenna we actually used some small, 50 ohm hardline (the RG-58-sized equivalent of "UT-141" PTFE coax) that was obtained on the surplus market, but RG-8x or even RG-58 would have been fine.
  • When the antenna was tuned up, it was mounted on a support placing it several feet/meters above the ground, an MFJ analyzer was connected to the far end of the coax (10-15 feet away) to minimize the effect of having a person too close to the antenna and affecting tuning.  For initial tuning, it should be mounted to the same type of mast as that which will be used for permanent mounting.  For the antennas at the WA7X beacon, black ABS water pipe has proven to be durable.  In the case of the 2 meter antenna, it was mounted using some PVC piping that seems to be holding up despite being out in the weather for well over a decade.
  • We then prepared two nonmetallic sticks - 5-6 feet long (1x1, wood, dowels, bamboo, etc.) and one of these had a piece of heavy wire to use as a shorting stub and the other had the balanced (200 ohm) side of the coaxial balun, also connected to 2 wires. The wires/balun were simply taped to the end of the stick to allow contact to be made.
  • Make sure that the copper pipe from which the matching section is made is clean and free of oxide - usually by sanding with fine-grit paper - to allow a reliable connection while finding a match and for ease of soldering.
  • At the position farthest from the antenna, the sliding shorting bar was placed while sliding connection to the balun was placed near it, on the "antenna" side of the shorting bar.
  • With the sticks, the two were slid around to achieve 50 ohm match.  While the two sliding portions are held in place, another person marks their position with a permanent marking pen on the antenna.  It is easier to move the connection on the balun back and forth while watching the VSWR while slowly moving the shorting bar back and forth, looking carefully for a match.
  • Once a preliminary match is found, the sliding shorting bar is replaced with a piece of heavy wire (#10-#14 AWG) that is wrapped around the pipe at the marked position. The other sliding bar (on the balun) is then re-checked for a good match, the shorting bar's position tweaked as necessary.
  • Once the position of the shorting bar has been established, the wire on the balun section is wrapped around the pipe.  This allows the wooden stick to be removed.  The positions of the two connections are then tweaked for best match.
  • The two connections are soldered in place, and the match re-checked.  If it is OK, connections are sealed and the match re-checked and adjusted as necessary.
  • In both Figure 1 and Figure 2 one can see small pieces of acetal (e.g. Delrin tm) plastic on the matching network - this material being chosen for its low RF loss characteristics and its durability to UV exposure.  Note that PTFE (a.k.a. Teflon tm) would have also worked well.  Some of the pieces (those at the far left edge of the matching section) are used for mechanical support. but the others are then for fine tuning:  The position of these pieces of dielectric slightly alter the tuning.  After the antenna was fully assembled, these were moved back and forth for the best match and secured in place with blobs of RTV (e.g. Silicone tm) sealant on both sides.

As can be seen in Figure 2 on both the 2 and 6 meter antennas, the attachment point of the balun is fairly close to the shorting bar.  The proximity of the coaxial cable balun to the match will affect tuning a bit, so it must be fixed into place before the final tuning is done.

Figure 2:
Annotated image showing the locations of the shorting bars, coax baluns, balun connections and the rain shields on
the 6 and 2 meter cycloid dipoles.  The matching network on the 6 meter antenna is longer than
necessary to allow its far end - which is electrically neutral - to be clamped to the mounting pipe and attached to
a ground wire for static discharge/lightning protection.  The connecting cables were secured with good-quality
electrical tape and black "zip" ties, which were also covered with electrical tape to protect them from UV.
Click on the image for a larger version
Note that for each antenna a "snow shield" (for rain, as well) was placed over the top of the matching section to minimize the effects of moisture, but the addition of this shield did change the tuning, as did the addition of RTV sealant on some of the connections, so final tuning must be done with such hardware and sealant in place.

The entire procedure is a lot easier if there are 2 or 3 people participating as it is pretty tricky for a single to hold two wires on sticks in place. If there is only one person, the shorting bar wire would be wrapped around the (clean!) pipe at a position correlating to about 0.4 wavelength on the pipe and the balun portion slid back and forth to see a "dip" in the VSWR, iteratively adjusting the shorting bar back and forth experimentally while sliding the connection from the balun to get the best match.

As I noted, it is possible that a 1/4 wave section would have been fine, but we just used the 1/2 section as there would be no doubt that it could be matched - and we wanted to minimize the hassle related soldering/unsoldering things as much as possible.  Importantly, this type of match - using the large pipes and "open wire" line - is very low loss compared to many other matching networks (e.g. those using small wound coils and discrete capacitors) and it contributes to the mechanical strength of the antenna itself.


This page stolen from ka7oei.blogspot.com


Tuesday, November 14, 2017

An alternate PA transistor for the QRP Labs Ultimate 3S beacon for more output power

The QRP Labs Ultimate 3S beacon (kit) is a reasonably-priced, compact and self-contained unit.  Despite its apparent simplicity and low price it is capable of transmitting in a variety of modes, such as WSPR, CW, Hellschrieber to name but a few on any amateur band from 2200 meters through the U.S. 222 MHz band.

Sort of.

Actually, it's not really that simple:  For operation on 160 through 10 meters, the construction is rather straightforward but for the higher (VHF) and lower (LF, MF) frequency bands, a few "mods" have to be made.  On the bands 160 through 10 meters the design of the circuitry means that at best, only a few hundred milliwatts of RF is possible with the parts supplied, the output power dropping off as one goes up in frequency.

The power amplifier:
Figure 1:
The front panel of my Ultimate 3S beacon, WSPRing away on
20 meters.  You can tell that I live in the U.S. by the position of
the "power" switch!

Click on the image for a larger version.

To minimize the cost, the power amplifier section (Q1-Q3) of the Ultimate 3S beacon uses BS170 N-channel low-power MOSFET transistors.  These devices are capable of dissipating about 1/3-2/3 of a watt each and there is room for three of these devices.  If an efficiency of about 50% can be obtained, it should be possible to safely get between 0.1 and 0.5 watts out of the beacon on the lower bands (e.g. 160-30 meters) - plenty for modes that allow very weak signals to be detected such as WSPR.

But, there is a problem.  The BS170 is not an RF transistor, but designed for low-power switching such as level conversion, turning on LEDs, small motors and relays.  At low frequencies - up to several MHz - it actually works quite well, capable of about a watt if three devices are installed, but by the time one gets to 10 meters it is, in this application, rather challenging to get more than about 100 milliwatts from the Ultimate 3 without a bit of tweaking.

One of the options the builder can choose is whether to wire the PA transistors for 5 volts, or connect it to a higher-voltage power supply.  In general, using a higher-voltage supply - say, 12-15 volts - will enable somewhat higher RF output power, but this also means that the same amount of bias current at 5 volts will result in higher power dissipation and finding the best value - without blowing up the transistors - is a bit of a delicate dance.

The problems:

The problems with this device at higher bands such 10/12 meters (and up) include:
  • Device capacitance.  There are a number of parasitic reactances involved - including the input and Miller capacitance.  All of these conspire to make it more difficult achieve a wide voltage swing and/or to turn the FET on and off quickly - something that needs to be done to amplify higher frequencies efficiently.=
  • The drive capability is rather limited.  The power amplifier section of the Ultimate 3 beacon is driven directly by the synthesizer which, for older units (mostly the non-"S" version) could be a DDS board, but the more recent versions use the Si5351 synthesizer chip which has a somewhat lower output level.  Neither of these devices produce enough output to "fully" drive the FET's gate.
The upshot is that while the rather simple amplifier circuit works pretty well at low frequencies, it drops off rapidly as one goes up.

One of the methods to deal with limited drive signals is to bias the transistor slightly.  Because it - like any similar FET - takes a volt or three to start turning on, biasing the transistor "on" slightly with a fixed DC voltage means that the limited RF drive signal doesn't have as "far to go" when it comes to driving the device.

Adding this bias works well - but only to a point:  Eventually, the transistor is conducting so much DC current that it is dissipating heat at/near its maximum rating and increasing the bias even more to further increase its effective gain is not an option.  One option is to add heat sinking (by gluing the transistors to a piece of aluminum or copper) to keep them cool, but this is of limited utility.

The Ultimate 3 beacon has the capability of using up to three of these transistors in parallel and while this can improve the power output at lower frequencies (maybe) the limited drive capability of the synthesizer - plus the fact that each transistor has its own capacitance - doesn't necessarily help.  One other factor often overlooked is that FETs are notoriously inconsistent in their DC characteristics:  Unless one goes through pains to match individual FETs - even devices from the same lot - when several are placed in parallel and biased, one is inevitably going to pull more drain current than the others.  This means than when several parallel devices are used, one or two are going to be doing most of the work and under stress while the other two (or one) will be doing comparatively little.

All of this would seem to be an argument to use a single, more capable amplifying transistor to obtain more output power.


The BS170 is quite popular in QRP transmitters because it is cheap, but it can be made to work "less badly" and the best way to do this is to strongly drive its gate with an RF signal.  Often, high-speed CMOS gates are used for this such as a 74AS04 or equivalent with multiple sections wired in parallel.  Doing this "brute force" drive technique can greatly improve the output capability of this otherwise low-frequency device and if done correctly, a DC bias is unneeded, saving a lot of hassle.  Unfortunately, the Ultimate 3 beacon doesn't have a device like this in its signal path, instead connecting the output of the synthesizer (more or less) directly to the gate of the output transistor(s), but one could hack the circuit and wire such a device into the circuit.

Another work-around would be the use of a transistor specifically designed for RF use.  While there are many such devices available, most are quite expensive or hard to find.

One such device is the RD16HHF1 made by Mitsubishi, but this transistor is becoming increasingly difficult to find.  Taking into account the fact that it may difficult to use the original (small!) holes for the BS170, using this device should work well - provided that it is operated within the capabilities afforded by the limited ability to dissipate heat.

The RD16HHF1 is also a favorite for counterfeiters that take an ordinary FET's die, put it in a package and label it as the real thing:  This fakery may work on lower bands, but it falls apart at higher bands for the same reasons that the BS170's efficacy drops off.  Some counterfeiters don't even bother to mount a fake die, instead taking an ordinary power FET and label it as an RD16HHF1:  Because the drain and source connections of the RD16HHF1 is backwards from "normal" FETs, a device like this will simply short out the power supply!  The only way to be absolutely sure that one has a genuine RD16HHF1 is to put it into a "component tester" - those inexpensive (<$20) devices that will identify practically anything - and see if its pin-out is correct and that its gate capacitance is in the 60-100pF area - and they try it in circuit.

The PD85004:

In perusing the catalogs I determined that a likely candidate device was the PD85004, made by ST Microdevices and available from a number of vendors such as Mouser Electronics.  This device, designed to operate from 13.8 volts, is rated to output several watts at 900 MHz, so it should surely be coaxed to work at HF, right?

This device is far more expensive than the original,  in single quantities costing about $3.25 each as opposed to about $0.50 each for the BS170 - but the expense isn't very onerous, and it is cheaper than a genuine RF16HHF1 - and it is also rated for operation at 13.8 volts, which happened to be the voltage that I used to power the final amplifier section of the beacon.
Figure 2:
The PD85004 mounted to an EvilBay SOT-89-4 carrier.  Short wire
leads go in holes 1, 2 and 3 to effectively turn it into a through-hole
device.  The heat sink was not yet added when this picture was
taken.  As described in the text, I eventually used a 10 ohms
resistor for the gate lead (position 1) to prevent circuit instability.
Click on the image for a slightly larger version.

One complication with the use of this device is that it is available only in a surface-mount package.  Fortunately, I had on hand some SOT-89-4 "carrier" boards (readily available on EvilBay - search for "SOT-89 adapter board") to which I soldered the device, effectively turning it into a leaded device that can be wired into the original FETs' board locations.  These boards cost anywhere from $0.03-$0.20 each, if you buy 10 or more - and that price often includes shipping!

To improve device dissipation a piece of copper flashing was carefully soldered to the tab of this device (which is the grounded source lead) after it was mounted to the carrier (see Figure 3.)  While the rated dissipation of this device is 6 watts, the mechanical layout of the Ultimate 3 beacon significantly limits the size of the heat sink as well as how much heat can be radiated/conducted to its surroundings.

Modifying the U3 for use with the PD85004:

Initially, I simply wired a PD85004 in place of a BS170 - but owing to the fact that the new device was designed to operate near 1 GHz - and that the layout of the U3 circuit board and interconnects didn't look to be particularly "VHF friendly" - so I expected that there could be some problems.  Before powering it up for the first time I turned the bias potentiometer all of the way down and pre-set my bench supply to current-limit at 500 milliamps - just in case I'd miswired something or I'd managed to turn the bias control all of the way up, instead.

Powering up the beacon and temporarily disabling transmit (easily done in WSPR mode by disconnecting the GPS antenna that is used for timing) I noted the current consumption - about 350 mA, much of that being the LCD's back light - and carefully adjusted the bias to cause a 100mA increase in current consumption.  With the antenna output of the beacon connected to a dummy load via a wattmeter I then reconnected the GPS antenna, readjusted the power supply current limiting and waited for the unit to come online and cycle through the various amateur bands while listening, in turn, to each frequency on a local receiver - using it as an oft-overlooked piece of useful test equipment that most amateur operators already own!

The result, not unexpected, was that I was able to get a power reading on each band, but on some bands - 160 through 40 meters - I heard a loud "hiss" +/- about 20 kHz from the transmit frequency instead of a CW note while the higher bands, 30 through 10 meters, sounded normal.  This just goes to show that at these frequencies this GHz-rated device may need some "taming" to prevent the apparent low-frequency instability and that a wattmeter alone is not necessarily useful for determining if an amplifier is working properly!

To tame the amplifier, I did several things:
  • I installed a 0.1uF between the wiper of the bias adjustment potentiometer R5 and ground.
  • I placed a 220 ohm resistor in parallel with R6 from the bias supply.  This, along with the added capacitor, helped "swamp" the drive signal and provide some lower impedance, low-frequency termination of the device's gate.
  • I replaced the wire lead on the SOT-89 carrier that provided the gate connection with a 10 ohm resistor - pin "1" on the device carrier board in the pictures.  This added resistance helps to break up effects of spurious reactances that can cause the transistor to behave badly in-circuit.
In testing with the dummy load I found that the amplifier section was now stable.

I then proceeded to carefully adjust the bias and watch the power meter.  Turning up the bias to several hundred milliamps I observed that I could get 2-3 watts of "clean" RF on on every band - but this much power was too great for the heat sink, but had an "infinite" heat sink been possible, I'm certain that I could have safely operated at this power level.  Monitoring the temperature of the heat sink I found that I could safely get about 1.5 watts out on 160-20 meters, dropping to about a watt on 10 meters, but erring on the side of caution I backed this off a bit to about 0.75 watts on 10 meters.
Figure 3:  
The installed PD85004 with heat sink in the Ultimate 3S beacon
for initial testing.  A piece of copper was soldered to the tab of the
transistor to allow it to dissipate a couple watts of heat.  Unfortunately,
there isn't really an easy way to make the heat sink capable of dissipating
significantly more heat without mechanically complicating things.
Not visible in this picture are the added 0.1uF capacitor between the wiper
of R5 (the pot visible to the left of the transistor) and ground
and the 220 ohm resistor in parallel with R6.
  Note that the connections from two of the three original
BS170 positions were used to wire this transistor into the circuit.  Behind
the transistor is the "OCXO" version of the synthesizer and I could have
soldered the heat sink to its case to improve dissipation, but this may risk
reducing the OCXO's own thermal and frequency stability.
This picture was taken before the 10 ohm resistor was put in series
with the transistor's gate to quell instability, replacing the wire lead
at position "1" on the carrier.
Again, the PA section of my beacon is wired to operate from 13.8 volts.
Click on the image for a larger version.

On the air testing:

My "main" HF antenna is a "lazy loop" of about 225 feet (approx. 70 meters) circumference at an average height of around 30 feet (about 10 meters) feed with 450 ohm window line with a 1:1 balun in the shack - designed to be connected to an antenna tuner.  Because I am not using the tuner with the U3 this means is my antenna it is not resonant (at 50 ohms) on any particular frequency, typically having a VSWR of greater than 5:1 on most bands.  While this may sound bad, the window line itself contributes negligible loss of its own and a reasonably-designed power amplifier should be able to tolerate such a mismatch.  I've run it this way for months with the BS170 finals without a problem and I've gotten reasonable signal reports.

When I connected the modified Ultimate 3S beacon to this antenna, everything worked fine - until I got to 40 meters, at which point I'd hear a loud "click" on the local receiver and the display would go blank.  Apparently, the bad (reactive) termination of the antenna caused the amplifier section to "take off" into some sort of mode of instability and somehow crash the beacon's processor.

Adding a "wee bit" of attenuation:

The work-around was to add an (approximately) 1.5dB resistive pad in series with output antenna connection.  Consisting of two 3.9 ohms resistors and a 220 ohm resistor in a "Tee" arrangement, this prevented the return loss as seen by the beacon from ever exceeding about 3dB, or a VSWR of about 6:1.  This little bit of padding reduced the transmit power only a fraction of an "S" unit, but with its added 3dB of return loss was sufficient to keep the amplifier stable on all of the bands.
Figure 4:
Typical "T" type resistive attenuator that can be useful for preventing
instability and/or damage to the final transistor in the event of a
poor match to 50 ohms.  These resistors were wired/mounted
at the RF output connector, after any low-pass filtering.
For Ra I used the standard values of 3.9 ohms and for Rb, 220 ohms,
resulting in approximately 1.5dB of attenuation.  For the 2.5 dB
attenuator on the WA7X beacon, Ra was 6.8 ohms and Rb was 180
ohms.  Neither of these sets of values are precise 50 ohm matches,
but they are more than "good enough"!

The addition of a bit of attenuation on a transmitter like this isn't necessarily a bad thing as it can offer a bit of protection - both in terms of VSWR and things like lightning strikes, offering both a DC discharge path and act as a bit of a "sponge" for induced spikes and excess power.  Even with this bit of attenuation I can safely coax about 1 watt of RF output on 160 through 20 meters, dropping to a bit over 0.5 watts on 10 meters - about 10dB better than I'd managed with a single BS170 on that band.  Because of the mismatch and commensurate losses I've currently set the WSPR beacon to report half this amount of power on some bands, but will increase it again when (?) I get around to putting up a multi-band matched antenna system.

The use of a resistive pad on the output of the transmitter had a precedent.  Upon installing another Ultimate 3S for the WA7X beacon at a remote cabin - this system operating exclusively on 10 meters - we discovered, the hard way, that the optional 5 watt amplifier (using an RD16HHF1)  didn't like it when the 10 meter vertical was temporarily detuned due to snow, causing a mismatch that resulted in the coincident failure of the output transistor.  In that case we added a 2.5 dB resistive pad (5 dB added return loss) to prevent the beacon from ever seeing worse than a 4.5:1 VSWR (even if the antenna connection were accidentally removed) and instead of 5 watts, the beacon is now operating at "2 watts" and is pretty "bullet-proof", reliability being very important for a remotely-controlled beacon at a remote location.

If that hadn't stopped the instability...

If the amplifier hadn't been adequately stabilized by the aforementioned modifications, there would have been two more things that I would have tried:
  • Add a 220-470 ohm resistor across T1, the output transformer, between V+ and the output transistor drain.  This resistor would help "Q-spoil" a low frequency resonance on the inductance of T1 that may cause similar oscillations.  This sort of instability is quite common, often due to the fact that RF devices can have tremendous gain at very low frequencies and the interaction with the rather large amount of inductance of the coupling transformer.
  • Add a series 1k resistor and 0.1uF capacitor between the output transistor drain and gate.  This degenerative feedback will also help quell spurious oscillations.
Ultimately, I hope to connect the beacon to an antenna (perhaps a trapped vertical) that is resonant on at least most of the bands on which the beacon operates, but in the mean time, this seems to be working out pretty well.

Final comments:

I started out this blog entry with the mention of bands above 10 meters, which naturally brings up the question:  Will this same modification work on 6 meters and higher?

The answer is yes, probably.

While I can imagine that it should be possible to obtain, perhaps, 0.25-0.5 watts on 6 meters with this same device and using similar techniques, going up much higher in frequency and getting some RF power will probably require a bit of modification as the board layouts and interconnects start to get a bit "iffy" at VHF and higher, requiring special care to avoid excessive harmonic content and other spurious signals.


Stolen from ka7oei.blogspot.com

Wednesday, November 1, 2017

A (semi)-typical suburban E-field whip receive system for the 630 and 2200 meter amateur bands

Even though the general availability of the 630 meter (472-479 kHz) and 2200 meter (135.7-137.8 kHz) bands to U.S. amateurs is a recent phenomenon, I've had interest in these frequency ranges for about as long as I can remember.  Back in the "old" days (the 1980s, for me) I would listen in these low-frequency ranges (10kHz to 530 kHz) using my modified Drake TR-7 which has an "LF input" on the back panel.

From the very beginning, I discovered a few things that did not work well for receiving these frequencies:
  • Simply connecting an end-fed random wire to the "Low Frequency" input.
  • Using my 40 meter dipole.
  • Anything that was indoors.
Attempts to do any of the above resulted in either no audible signals other than a racket of power mains "buzz" that would drown out anything that I could hope to hear.  I quickly realized that there were a few signals that I could hear without too much trouble - mostly the very high-power VLF transmitters between 17 and 30 kHz and WWVB at 60 kHz, which is only few hundred miles/km away - and I knew that unless I could hear those signals really well that there would be little hope of hearing anything that was actually weak.

The "discoveries":

Figure 1:
The LF-400B active e-field whip on my
roof.  The antenna is about 5 feet (1.5
meters) above the roof, mounted to a vent
pipe.  The red ground wire connected to
the coaxial cable's shield at the bottom
of the antenna can be seen along with
Click on the image for a larger version.
Even as a teenager with limited experience and knowledge in such things I realized that at such long wavelengths even a rather long piece of wire as a receive antenna would be akin to putting a paper clip in the antenna connection of an HF rig and expect to "hear the world" - but I also knew that it was possible to hear low frequencies quite well on a very short antenna:  The short whip on my car could hear the entire AM broadcast band really well - so it was possible if done correctly.

These realizations told me several things:
  • I would probably have to match the "short" antenna to the receiver input to be able to hear anything.  I determined that this could be done with a series inductor or some sort of high-impedance amplifier - or a combination of both.
  • When in a car, I could be well-away from interference sources - such as power lines and noisy appliances - and could hear weak AM stations.  Somehow I had to keep the interference from things in the house from finding their way into my receiver.
Rummaging around in my junk box I found a large, variable inductor - probably from a scrapped TV - that I placed in series with my wire antenna and receiver - and over a limited frequency range (dictated by the adjustment range of this inductor) I noticed a dramatic improvement in the signal strength at about the frequencies that the combination of the coil and antenna provided a semblance of matching - although the noise was still substantial.

The next "breakthrough" was to wind a simple 1:1 transformer on a chunk of ferrite - probably the flyback transformer core of an old TV -  that allowed only magnetic coupling between the radio and its chassis, and the antenna and a connection that went directly to my kludgy system of buried ground rods.  By doing this, the "noisy" ground of my receiver - which was connected throughout the house with its noisy devices - was no longer referenced to the antenna.  Because the antenna must have a "ground" of some sort to "push" against I knew that if that "ground" was the radio itself, which was connected to the noisy house wiring, that this noise would, in effect, appear on the wire antenna.  This transformer effectively decoupled the two, using, instead, the comparatively "pristine" ground rod for the antenna to "push" against.  (For a depiction of this method see the external link to a paper by DL1DBC at the bottom of this page.)

Between the above two tricks an entirely new world opened up as I could now hear the (now defunct) Omega transmitters between about 10 and 14 kHz and a myriad of "NDBs" (non-directional beacons) and similar signals in the range from 190 through just below 530 kHz.  To be sure, I had to do most of my listening at night when TVs and lights were turned off, but that's when most of these frequencies propagated best, anyway!

The "LowFER" band:

Somewhere around this time I learned of the so-called 1750 meter "LowFER" band - a spectral slice from 160 through 190 kHz where legal, unlicensed operation (according to FCC §15.217 - read more here) could occur with some very strict limitations (e.g. and antenna that was, at most, 15 meters "long" and a maximum of 1 watt of input power.) but the challenge of both transmitting a usable signal with these limitations and receiving it via conventional techniques (e.g. CW) had its appeal.

It was at about this time - in the mid 1980s - that I purchased an LF Engineering LF-400B - a commercially-available active E-field whip antenna that seemed to have decent reviews in the various longwave-related newsletters to which I then subscribed.  This antenna, with a built-in amplifier and a strong low-pass filter to remove signals above 500 kHz, was much more convenient than trying to string a long piece of wire and matching it as it was rated from "3 kHz to 500 kHz".  One slight disadvantage of this - or any active antenna - is that it needs power, supplied in this case by a "power inserter" that ran from an external power supply or a pair of contained 9 volt batteries.

Being an E-field whip antenna it was still sensitive to the direct radiation of interference from the household and neighborhood wiring and appliances, but provided that I located it away from the house and "decoupled" its cable by winding as many turns as could fit on the core of a flyback transformer from a scrapped TV and grounding the shield at the antenna, it seemed to hear the background static very well - and if I could hear the background noise, there was hope that I could hear the weak signals buried within.

It was during this time period that I actively listened on the LowFER band, managing to hear a number of stations that were 200-700 miles (about 300-1100 km) away and, on one winter evening, hearing a station halfway across the continent - about 2000 miles (3200 km) away.   I also set up my own LowFER beacon that, although very modest, was occasionally heard, on CW, up to 700 miles (1100 km) away.

Another antenna to consider for MF/LF/VLF reception is a shielded H-field loop.  By its nature, it is less-sensitive to nearby E-field energy - often that which emanates from electrical devices' interference radiating from wiring.
Another advantage of a loop is that it has a "figure-8" pattern with two nulls, allowing the possibility of rotating it such that one of these nulls is oriented toward an interference source.  The obvious disadvantage is that a loop should have provisions for rotation to steer it into the null for the worst interference - or take care of those instances where the desired station happens to be in the direction of the null.
Shielded loops are available and they can be constructed fairly easily, typically using a piece of coaxial cable.  Unless they are rather large and/or actively tuned to the receive frequency, they - like a short E-field whip - must have an amplifier that is externally powered.

Fast forward to the 21st century:

As it happens, I still have the L-400B and it has been outside, on a roof, for most of the time since the mid 1980s.  Other than having to repair it a time or two (usually due to condensation and related corrosion) it still works as well as it ever did.  While I had not been as active on LF as I once was, I've been maintaining that receive antenna and with the recent availability of the 630 and 2200 meter bands, interest has been rekindled.

To this end, I decided to document my receive antenna installation, showing what "works for me."

The antenna on the roof:

I will admit to a luxury that most others will not have:  My house has a metal roof.

Figure 2:
A close-up of the coax choke at the antenna.  This
choke consists of 10 turns wound on a large
ferrite bar.  The coax used is solid-dielectric RG-58.
The use of a solid dielectric rather than a foam
dielectric - such as that found in RG-6 - allowed
a very tight radius winding without worrying much
about the center conductor "migrating" and shorting
to the shield.  A cable like RG-174 would have also
been usable, allowing a tight radius and more turns.
At these frequencies, the loss of the coaxial cable is
Click on the image for a larger version.
The metal roof not only acts as an excellent ground plane, but it is also an effective barrier between what is "inside" my house and the "outside world".  This means that at VLF and LF frequencies, things in my house that generate noise (light dimmers, switching power supplies, TVs) are fairly effectively isolated from this antenna on the roof - at least in terms of direct radiation of energy from these devices.

If you are not "blessed" with a metal roof on your house - but you are willing to go through a bit of hassle - you could lay down a suitable ground plane:  Many people have been known to put down a layer of chicken wire on the roof or an interconnected grid of wires to act as an effective shield.  Practically speaking, it need not cover the entire roof, but if the radius of this plane is 1-2 times the height of the antenna over the roof, it will probably have reasonable effectiveness.

Somewhere this plane must be grounded and it is best that this is done via its very own ground system - which could be as simple as a ground rod - which is preferred over tying into the house's "noisy" electrical ground.

As can be seen from the picture in Figure 1 the whip antenna is mounted to a vent pipe at a height of approximately 5 feet (1.5 meters) above the roof - which happened to be the length of the piece of aluminum that I'd found to mount the antenna.  When experimenting with mounting this antenna I found that if I placed it just above the metal roof, it was very quiet and relatively insensitive - but much of that was due to the fact that the very E fields to which the antenna is sensitive decrease significantly with proximity to "ground".  By raising the antenna above the roof the signals increase very dramatically, but still seemed to be within the "cone of silence" afforded by the metal roof.

Decoupling the coaxial cable at the receive antenna:

At the time that I bought the LF-400B antenna it was offered only with a permanently attached RG-174 feedline, but after about a year of use, often hauling it into the wild to listen, away from the city, the coaxial cable fatigued and broke, so I carefully disassembled it and installed a BNC connector (later versions of this antenna have a choice of connectors as an option.)  This modification allowed me to connect a ground directly to the bottom of the antenna.
Figure 3:
A block diagram of the antenna and receive system showing the grounding and coax chokes.
Note that the "roof ground" - which is, in my case, the metal roof itself, but it could be a grid of wire or fencing material laid on the roof and is grounded elsewhere - is connected directly at the shield of the e-field whip itself, "before" the coaxial choke.  At the ground level, in close proximity to the building entry is another connection to a "clean" local ground such as several ground rods and/or some buried ground radials.
The "Inside coax choke" has the most inductance and does most of the isolating of common-mode noise currents that could otherwise "light up" the antenna with electrical noise that would be conducted from the radio system's ground connection to the power mains.  The DC power inserter puts DC on the coaxial cable to the antenna - but not on the coax to the receiver - to provide power.
Click on the image for a larger version.

As mentioned earlier, one of the "tricks" to a quiet E-field antenna is to prevent electrical noise from being conducted from the receiver and "lighting up the ground" of the antenna itself - a problem that is arguably worse than the antenna itself picking up noise, directly.  One of the better ways to to do this is to "decouple" the coaxial cable between the antenna and receiver using a large amount of inductance on the feedline - and I chose to do this several ways.

As can be seen from figure 1 there is a (red) wire connected directly to the antenna's connector that, in turn, connects to the "local ground" - that is, the metal roof itself.  By doing this, the "ground" of the antenna and the roof are at the same RF potential and the interception of "local" interference by the antenna is reduced.

Also visible in figure 1 - and in more detail in figure 2 - is an inductor in the form of a portion of the connecting coaxial cable being wound around a large ferrite rod from a discarded AM radio.  The location of this inductor places it between the antenna and the receiver and its inductance adds common mode impedance to signals that would be conducted along the coaxial cable, but will not affect the desired signals within the cable itself.  A better choke for this location would be like that depicted in Figure 4 (and described below) as it has higher effective resistance at the frequencies of interest, but since I'd already installed this one, I left it in place.

Figure 4 shows the other end of the cable just after enters the house.  Just as it enters through the window there is another BNC connector, and connected to the shield at that point is a wire that goes directly to a grounding system that is located immediately outside the window.  Between this grounding point and the inside of the house where the connection to the radio is made the coaxial cable is wound around a much more substantial choke - this one consisting of as many turns of the RG-58 coaxial cable as will fit on a TV flyback transformer ferrite core that was scavanged from a discarded CRT TV or computer monitor.  The details of the locations of these chokes and the grounding points is detailed in Figure 3.

It is this second "inside" choke that does most of the work:  Consisting of about 20 turns, it has a measured inductance of about 15 millihenries.  In running the math we can see that this large amount of inductance is what is required to effectively isolate the coax at LF and VLF frequencies, as in:

Where inductive reactance is calculated using the equation:
Z = 2 * Pi * F * L
Z = Reactance in ohms
F = Frequency in Hz
L = Inductance in Henries
Because we are dealing with milliHenries and kHz, the "10s" parts cancel out, so:

At 500 kHz:

500 kHz * 15 milliHenries * 6.28 = 47100 ohms

Because this is a linear equation, we can then re-run the numbers which tells us that at 50 kHz, the reactance is 4710 ohms and that at 5 kHz it would be 471 ohms.

What this shows us is that even at very low (VLF) frequencies, our rather substantial inductance is still effective, so it will work nicely at both 630 and 2200 meters - and everything in between!

Figure 4:
The indoor coax choke consists of 20 turns of  RG-58 wound on a TV flyback transformer core.  This choke, with a measured inductance of about 15 milliHenries, provides excellent isolation even down below 10 kHz.  If a flyback transformer core cannot be found, a suitable choke can be wound on a high-permeability ferrite core using smaller (e.g. RG-174) coaxial cable as described below.  Note that to be effective at these frequencies this choke really does need to have at least several milliHenries of inductance!
Click on the image for a larger version.
Obtaining the inductance:

While "current-mode" 1:1 baluns that isolate the feedline in the manner we desire are readily available, unless they were specifically designed for LF and VLF use they do not have enough reactance to operate effectively at these low frequencies!  What this means is that unless a suitable product is offered by one of these companies that is has been designed for LF and VLF use, they will not work well!  This means is that you will probably need to construct your own coaxial choke.

Using flyback transformer cores:

Many years ago it was pretty easy to scavange flyback transformer cores from old CRT-based TVs or computer monitors, but these are getting harder to find - but it is a good thing since these transformers were part of the very circuit that caused a lot of interference at VLF and LF frequencies!  Once one manages to get the core out of an old flyback transformer in the first place (sometimes a trick in and of itself!) the fact that these cores are in two pieces makes it easy to wind the coaxial cable over one half and then assemble it.  When I come across a flyback transformer, I often resort to putting it in a toaster oven and heating it so that the glue softens.  Often, the core breaks - but ferrite typically breaks very cleanly and the two pieces can be rejoined using a drop of cyanoacrylate (e.g. "super") glue with little change in performance.

If a flyback transformer core is not available, what can be used, instead?

Using high-permeability toroidal cores:

While not as convenient as a flyback transformer core - which can be disassembled during winding - a ferrite toroidal core can be used, instead.  To maximize the number of turns, smaller coax such as RG-174 would be used and the connectors installed/connected after winding was complete.

Take, for example, a common ferrite material designed for low frequencies - "Mix 75" (sometimes called "Mix J") with a typical permeability of about 5000.  A reasonably large toroidal core would be the FT-240 (the complete part number would be either "FT-240-75" or "F240-75").  Note that the ferrite mixes that one would normally use for things like HF baluns aren't ideal for this purpose as they have lower permeability.

Extrapolating from a data sheet and rewriting the equation we can see that if we can manage to wind 30 turns on this particular toroidal core, we can expect:

L = Al * (T/1000)2

L = Inductance in mH
T = Turns
Al = mH per 1000 turns from the spec. sheet - 6850 for an FT-240-75


6850 * (30/1000)2 = 6.165mH

Clearly, this is a bit less than half as much as I'd measured on my discarded TV flyback, but if we use the equation above we still get 194 ohms at 5 kHz and over 5 kohms at 2200 meters - a respectable amount of reactance!  Using this size of core (an inside diameter of 1.4 inches/3.5cm) it is likely that more than 30 turns of RG-174 could be wound on it - and if you make this type of core, by all means, put as many turns on at as you can!

Unfortunately, the "Mix 75" toroids are not as easy to find as typical toroids designed for higher (HF) frequencies and if we use a more common type such as Mix 31 the result will be between a quarter and a fifth of the inductance for the same number of turns whereas "Mix 77" will, for the same number of turns, yield about 1/3 of the inductance as Mix 75, but this would still imply between 1 and 2 kohms at 2200 meters - still quite good.

Where does one get this sort of toroid?   Toroids can be found at a number of places, including:
  • Palomar Engineers (link) 
  • Amidon Associates (link)
  • Another distributor of some of these devices is the web site kf7p.com - link.
Again, while "Mix 75" is preferred, "Mix 77" is the second choice - and cores may be stacked to increase the inductance for a given number of turns.

Another possibility - Common-mode chokes:

While a coaxial-based choke is preferred, there are other devices - possibly in your junk box - that may be suitable:  A common-mode choke used for power supply filtering.  The best place to find these is from scrapped switching power supplies - such as those used in computers.

Figure 5:
An assortment of power line filtering chokes and devices.  In the upper-left
is a self-contained AC line filter, but it is not suitable for this purpose as it
is designed to block all RF - both differential and common-mode.  All of
the other devices are dual-winding common-mode chokes that allow
differential currents to pass, but will block common-mode currents - but
not all of these devices are suitable for our purpose - see text.
Click on the image for a larger version. 
Figure 5 shows an assortment of typical devices - but not all of them are suitable.   As noted in the caption, the self-contained power line filter (upper-left) blocks all RF and wouldn't work, but the other devices allow differential currents to flow while blocking common-mode currents - which is what we want.

In order for these devices to be suitable for our purpose, they need to have:
  • Adequate inductance.  As we noted above, we need milliHenries of inductance to effectively choke out interference at LF and VLF frequencies.  The smaller toroidal chokes shown - typically those wound on toroidal cores - have hundreds of microHenries of inductance which may be suitable at 630 meters, but could be marginal at 2200 meters.  For example, a choke with 100 microHenries per winding will offer about 295 ohms of reactance at 630 meters, but only 86 ohms at 2200 meters.  Because we want as much reactance as possible - at least in the many hundreds of ohms - we would hope to do better!
  • Good balance.  All of these chokes consists of two identical windings and the idea is that if a common mode signal appears across both windings, they will be suppressed.  If, however, the two windings are not identical, this suppression will be incomplete.  It is likely that the "transformer-looking" chokes (e.g. those that do NOT look like toroids) will have reasonable suppression at 2200 meters - and maybe even 630 meters - but as one goes up in frequency even more, the imbalance will grow.
  • Low loss to differential signals.  The reason that we can pass a signal through a coaxial cable wound on a large piece of ferrite without affecting the signal being carried by that cable is that the coaxial cable, by its very nature, is fairly low loss to the signals carried within where the signal on the inside conductor of the coax is precisely equal and opposite to that carried on the shield.  If one has separate windings, each carrying an equal and opposite signals, imperfections in these two windings - sometimes the same as those that cause imbalance - can cause degradation of those signals.  As one goes up in frequency these ferrite cores - which are formulated to block low frequencies - can start to get lossy - and this doesn't include the self-capacitance of the windings which can cause other things to happen, such as strange resonances or coupling.  In other words, they may work find at low frequencies, but "fall apart" at higher frequencies such as 160 meters (1.8 MHz) and up.
Figure 6: 
An example of how a bifilar (or similar) choke would be
connected to a coaxial cable.
The diagram above depicts how the two windings would be
connected, keeping straight which is the "center", and that
which is the shield of the coaxial cable.  The dots indicate "phasing" -
that is, same ends of the two windings connect to the antenna side and
the other ends connect to the receiver side.
Click on the image for a larger version.
In short, the suitability these devices for our purpose is best determined experimentally.

How it is would be connected:

Figure 6 shows how such a device would be connected to coaxial connectors.  Note that the winding for the shield on one side of the choke connects to the same shield on the other side.  In theory, this wouldn't matter at RF, but because we may need to conduct DC to power the active antenna, we would also need to preserve the polarity.

Not also that both sides of the input and output coaxes connect to the same "side" of the dual winding choke as indicated by the dots - in other words, the two windings are in phase with each other:  Were either one of the windings (ground or center conductor) "flipped", this choke would do exactly opposite that which we desire - that is, the signal on the coax would be blocked, leaving only noise!

For an inductor such as that depicted in Figures 5 and 6 that is not wound with coax, it doesn't matter which side is the shield and which is the center - just as long as the windings are "shield-to-shield" and "center-to-center".

Another example of feedline choking and grounding:

Figure 7:
RG-174 coaxial cable wound on a TV flyback core.  About
55 turns fit on this core yielding a measured inductance
of a bit over 1.5 milliHenries.  If this core
had equal gapping in both of its "legs" (see text)
the inductance would have been higher.
Click on the image for a larger version.
Figure 7 shows another example of how an E-field antenna's feedline (not mine) was isolated.

In this case - which just happened to be another LF-400B - RG-174 coaxial cable was permanently attached to the antenna.  This cable was cut, leaving about 20 feet (approx. 6 meters) of it still attached to the antenna and, leaving a "service" loop of about 1.5 feet (35cm) the remainder was wrapped on the flyback core of a discarded computer monitor.

It is worth noting that these flyback cores are usually "gapped" - that is, a small - usually plastic - insulator is placed between the two halves of a core to prevent it's being saturated.  On some of these cores there are two equal gaps - one on each of the two mating surfaces and if these are removed, the two ferrite surfaces mate closely.  In the case of the core in Figure 7, only one of these mating surfaces had a gap, meaning that one side mated closely while there was a gap on the other side.  This has the inevitable result of reducing the total inductance of the core.

In the case of the flyback pictured in Figure 7 the plastic gapping material was carefully retained and a very thin layer of epoxy was put on the two sets of mating surfaces and the metal bail holding the two together was reinstalled, the cores being worked back-and-forth to squeeze out extra epoxy.  Once this was done epoxy was applied to the wire bail itself to keep it in place.  After the epoxy was allowed to cure, the remaining RG-174 coaxial cable was wound on it, filling it up.

Figure 8:
A wire attached to the shield to permit grounding, necessary
because the coaxial cable was permanently attached to
 the antenna, preventing a connection from being made
at then antenna, between it and the choke.  See text.
Note that the ground wire emerges from the "downstream"
side of the coax.  When installed, this ground wire will
face down to reduce the probability of moisture ingress.

Click on the image for a larger version.
Figure 7 shows this core mounted in a plastic "pull" box intended for non-metallic electrical conduit.  On one end of the box is mounted an "F" connector to which the end of the RG-174 is soldered while the other end - connected to the whip antenna - emerges through a plug:  If you do it this way you will surely want to pull the coax through the plug before winding it on the core!  After it was assembled, it occurred to us that we should have put the "F" connector on the same side as the plug so that they could both be faced downwards.  Between the receiver and the choke box, ordinary RG-6 TV coax will work:  The impedance mismatch/loss is unimportant at this frequency, in this application.

Figure 8 shows a bit of detail about the grounding of the antenna.  This particular antenna has a permanently attached cable and the owner didn't wish to modify the antenna to add a coaxial connector to it.  That which follows was done before the coaxial cable was wound on the ferrite core.

The ground connection needed to be made directly to the cable's shield and this was done by carefully baring a bit of the shield by removing a small amount of the outer jacket and then using a hot soldering iron to quickly make the connection without melting the inner dielectric - a bit of a trick to do if one isn't skilled in the art of soldering!  To make this connection weather proof the connection was covered with a thin layer of thermoset (e.g. "hot melt") glue and a small piece of heat shrink tubing was slid over the joint and shrunk.  Over the top of this a thin layer of RTV ("Silicone") sealant was spread over the entirety of the connection and another, slightly longer piece of heat shrink tubing was installed and shrunk - and then another thin layer of RTV and slightly longer heat shrink tubing.   

While this sounds like overkill, it should prevent moisture from finding its way in between the jacket of the coax and the tubing.  Finally, this connection should be oriented at the time of installation such that water runs away from it - which is to say, the part with the wire coming out from underneath the tubing should be facing down.

More information about interference reduction:

While the above techniques will go a long way to reduce the amount of noise picked up by an E-field antenna - and, to a degree, any antenna - it is too-often the case that there will be some device that simply radiates a lot of noise.  While at HF frequencies and higher it is possible to reduce this noise with the application of large ferrite devices on cables, power cords, etc. this tactic simply does not work well at VLF/LF/MF frequencies because it takes so much reactance (inductance) to introduce enough effective resistance in the wire conveying this noise and a "snap-on" choke simply cannot do this.  Even if a device contains "good" noise suppressing components (not all do!) they simply may not be very effective at VLF/LF/MF frequencies.

If you are interested in listening on the LF and MF amateur bands, the necessary first steps are outlined above:  Do what is necessary to prevent noise from being conducted out, onto the antenna in the first place.

Once that is done, you may need to "seek and destroy" devices that are particularly egregious when it comes to generation RF "grunge" - and the typical suspects are switching power supplies, light dimmers and some brands of LED lights.  Plasma TVs are notoriously bad interference generators, but since they are no longer being made, their contribution to the miasma of QRM is slowly decreasing as they die off.

The best way to find noise that you can do something about is to power the receiver from a battery (NOT including an inverter!) and turn off all of the power to the house - including shutting down any UPSs that you might have.  If the noise decreases or goes away, turn on one circuit at a time until it returns and upon finding the circuit, isolate the specific device that causes the problem.  If the noise is just the same with your power off as it is on, there may be a noisy power line nearby and/or a neighbor may have a noisy device - and how you deal with those two entities is up to you!

If you find a device (or devices) that generate lots of interference, they might either be replaced with "quieter" ones or modified to be quiet.  Unfortunately, the latter can be a challenge and the links below include techniques for doing this.  If your goal is interference reduction at VLF/LF/MF - and you are constructing better filtering - remember that the higher-inductance chokes will be best!

How well does my receive antenna system work?

In the late evening and overnight, I can easily hear the "band noise" - that is, the sounds of the ionosphere and propagated storm static.  During the day time the noise level is typically lower as it seems as though propagated noise from a wide geographical area is suppressed somewhat - possibly by the formation of the ionospheric E-layer.  During the "busy" hours - particularly from, say, 5 to 11 PM, there can be a bit of interference from other peoples' TVs, appliances and whatnot, but it is usually not severe enough to completely quash reception.

In my ham shack I have some track lighting over the workbench that is equipped with LED floodlights and is controlled by a light dimmer.  While I do not "hear" the LED's switching power supplies, I do get a significant "buzz" on 2200 meters from the dimmer - but I don't hear it on 630 meters.  The work-around for this is to use a smaller work light near the workbench - both of them being fluorescent - one having an iron ballast and the other electronic - but neither of them causing detectable interference on either 630 or 2200 meters.

For the past 5 years or so there have been a number of Canadian amateur stations (who have had access to the frequencies around 630 and 2200 meters for a while) plus some U.S. based "experimental" stations that have also operated on a number of other frequencies and in this time, I've been able to "receive" these stations which are typically using a digital mode like WSPR or a more analog-like mode like QRSS (slow-speed Morse code) - both typically being detected by computer.  The operational frequencies of these stations has varied from above 500 kHz to below 30 kHz, depending on the authorized frequencies of the various experimental stations and I've generally been able receive such signals including a number of stations operating in the 470-500 kHz range across the U.S. and an experimental station operating near 29 kHz (yes, 29 kHz!) from New York state to my QTH in Utah - a distance of about 2000 miles (approx. 3200km).

In the relatively short time since U.S. amateurs have been allowed to operate on the 630 and 2200 meter bands I've heard several stations on both bands - some well enough to have copied using Morse code via ear and, possibly, even SSB voice.  As the northern hemisphere descends into winter - and as more amateurs receive authorization and put their systems on the air - I expect to hear even more stations.

One device in my arsenal is a "Line Synchronous Noise Blanker" - that is, a device that will mute the antenna signal when an interfering pulse - which is usually in sync with the power mains - comes in.  This devices is adjusted manually and can go a long way to knocking out this type of noise.  This device is described on this page:  A Line-Synchronous Noise Blanker for VLF/LF/MF use - link.

Links to other articles about power supply noise reduction:
Other information about the use of active antennas at VLF, LF and MF frequencies:

  • Discussion from the DL1DBC web site about active antennas, including their operation and installation - link.
  • Construction and installation of a PA0RDT whip by VK6YSF - link.
  • Discussion of E-field whip antennas by PA3FWM - link.

Final comments:

The L-400B still seems to be available - at about twice the price as it was when I bought mine in 1987.  The page with information on this and similar products may be found here - link.

In addition to the L-400B, there are now other active whips, including the AMRAD active whip, the PA0RDT "Mini-whip" and variants on those designs which may or may not include a low-pass filter to remove mediumwave signals.  All of these are reported to work well, but be aware that some receivers have difficulty dealing with signal from strong, local AM broadcast transmitters.


This page stolen from ka7oei.blogspot.com

Tuesday, October 17, 2017

A 10 MHz OCXO (Oven-controlled Crystal Oscillator)

Figure 1:
The 10 MHz OCXO (lower right) in use with my homebrew
24 GHz transverter.  At 24 GHz, the oven provides excellent frequency
stability, suitable for SSB or even digital modes, while providing a
frequency uncertainty of a few hundred Hz at most.
Click on the image for a larger version.
Why a frequency reference?

When operating on the microwave amateur radio bands, narrowband modes (such as SSB or CW) are often used to maximize the link margin - that is, to be able to talk when signals are weak - and when we use microwave frequencies and narrowband modes such as SSB or CW one must maintain pretty good frequency stability and accuracy:
  • Stability is important as a drift of even a few hundred Hz at the operating frequency (in the GHz range!) can affect intelligibility of voice - or, if CW is being used for weak-signal work, such drifting can move the received signal outside the receiver's passband filter!  Having to "chase" the frequency around is not only distracting, but it complicates being able to communicate in the first place.
  • Accuracy is also important because it is important that both parties be confident that their operating frequencies are reasonably close.  If a contact is arranged beforehand it is vital that both parties be able to find each other simply by knowing the intended frequency of communication and as long as the two parties are within several hundred Hz of each other it is likely that they will be able to find each other if the path "works" in the first place.  If the error was on the order of several kHz, "hunting" would be required to find the signal and if those signals are weak, they may be missed entirely.
Because achieving such stability and accuracy requires some effort, it is more convenient if our gear is constructed such that it can use a common, external frequency reference and lock to it.  In that way, we need only have one "master" reference rather than several individual references.

Figure 2:
The 10 MHz Isotemp 134-10 OCXO - one of many similar units that
often show up on EvilBay.  A 200uF, 16 volt capacitor is soldered
directly to the supply terminals of the OCXO to provide low-impedance
filtering of any noise that might appear on it - any value from 2000 and
up (to several thousand uF) would be just fine.  The green device is a 10-turn
trimmer potentiometer soldered directly to the OCXO's pins.  This
potentiometer is used to adjust the tuning voltage to precisely set the
frequency and locating it at the OCXO practically eliminates the possibility
of external noise pick-up on the tuning lines and the possibility of the I*R
drop on the wires causing a slight tuning shift as the oven power changes.
The OCXO is mounted in the case using rubber/metal shock mounts with "blobs"
of RTV (silicone) on the sides that prevent it from hitting the inside of the box
should the unit be accidentally dropped.
The corners/edges of the OCXO could be mounted in some stiff foam,
instead - but it should not be thermally insulated by this foam unless you have
demonstrated to yourself that doing so will not reduce the oven's stability.
Click on the image for a larger version.
Having one common frequency reference can also be convenient if one is operating portable using battery power since it can mean that one doesn't need to keep all of those individual pieces of gear "warmed up" all of the time to maintain stability.  If a particular piece of gear can accept an external 10 MHz input, this would allow one to turn on that gear (and drain battery power) only when it is needed.

At this point I might mention that Rubidium frequency references (such as one described here) are also readily available in the surplus market as well that provide at least an order or magnitude greater accuracy and stability and warm up in less time than the crystal reference, so why not always use a Rubidium reference instead of a crystal-based one?  The crystal-based unit is cheaper, easier to package and consumes significantly less power than a Rubidium reference, and the stability/accuracy of a good-quality crystal-based reference is more than "good enough" through at least 24 GHz.  When I go out in the field to do portable microwave work I'll often power up the OCXO after putting it in the car knowing that by the time that I get to my destination and set up, it will be warm and on-frequency.  (To be sure, I bring a Rubidium reference as a "backup"!)

About this frequency reference:

The oscillator:

The goal for this project was to have a "reasonably stable and accurate" reference:  Based on an Isotemp OCXO 134-10 this particular unit has a rated stability of about +/-1.0x10-8 (+/-1 Hz at 100 MHz) or better after it has warmed up for a while with short term variations approaching +/-1.0x10-10 (+/-1 Hz at 10 GHz).  In-field observations appear to confirm this stability with tests having shown that this unit seems to be able to hold the 24 GHz local oscillator to within 500 Hz or better with no obvious frequency "warble" once it has had 15-20 minutes or so to warm up -  and it seems to be fairly stable across a range ambient temperatures from "hot" to "below freezing."  The Isotemp unit - and others like it - are readily available on both the new and surplus markets, available via EvilBay and similar and other than having different voltage and stability specifications, they, too, can be integrated into a stand-alone project such as this.

The oven module itself is rated to operate from 13 volts, +/- 2 volts, implying a minimum of 11.0 volts.  Even though testing indicated that it seemed to be "happy" with a supply voltage as low as 9.8 volts or so, it was decided to adhere to the published specifications and in looking around I noticed that most readily-available low-dropout regulators (and those that I had onhand) were not specified to handle the maximum "cold" current of this oven - about 800 mA - so I had to "roll my own" 11 volt "zero-dropout" regulator.  More on alternative regulators, below.
Figure 3:
The inside of the enclosure containing the OCXO, regulator and driver.
On the left is the shock-mounted OCXO while the circuit on the perfboard
is the "zero drop-out" regulator and the 10 MHz distribution amplifier.
The P-channel FET pass transistor can be seen along the top edge of
the die-cast enclosure, bolted to it to dissipate any heat while along
the right edge, inside the enclosure is a piece of glass-epoxy circuit
board material to provide a solid, solderable ground plane for the
distribution outputs and the DC input filtering.

A "zero-dropout" regulator:

Why regulate?  I noted in testing that slight variations of supply voltage (a few hundred millivolts) would cause measurable disturbances in the oscillator frequency due to the changes of the power applied to the heater, taking several minutes to again reach (thermal?) equilibrium.  Since battery operation was anticipated, it is expected that the supply voltage would change frequently between periods of transmit and receive - as well as due to normal battery discharge.  Because I had chosen to use an OCXO that required (at least) 11.0 volts to be run from a "12 volt" lead-acid battery, I needed a circuit that would reliably produce that 11.0 volts even when the battery voltage dipped below 11.5 volts - as it could during heavy transmit loads and the end of a power cable with the battery near the end of its charge.

Referring to the schematic U101, a standard 5 volt regulator (the lower-power 78L05 is a good choice) provides a stable voltage reference for U103, a 741 op amp, which is used as an error amplifier.  A 7805 was chosen as it is readily-available but a Zener diode and resistor could have been chosen:  If a Zener is used, a 5.6-6.2 volt unit is recommended with 2-5 milliamps of bias as this voltage range offers good temperature stability.

If the output voltage is too low, the voltage on pin 3 (the non-inverting input) drops, along with pin 6, the op amp's output which turns on Q103, a P-Channel power MOSFET by pulling it's gate toward ground, which increases the voltage and once the voltage on the wiper of R119 reaches 5 volts - that of the reference, which is applied to pin 2, the non-inverting input - the circuit comes to equilibrium.  A P-Channel FET (a slightly less-common device than an N-channel) was used because it takes 3-5 volts of drain-gate voltage to turn on a FET and it would have been necessary to have at least  3-5 volts above the power supply (about 16 volts) to bias the gate "on" if an N-Channel FET were used whereas we can pull the gate voltage "down" from the supply voltage with a P-channel device.  Furthermore, with the use of a P-Channel power MOSFET the dropout voltage of the regulator is essentially limited to the channel resistance of the that FET.  In theory a PNP (possibly a complimentary pair arrangement) could be used instead if one can tolerate closer to a volt of dropout, but the FET was chosen to minimize the dropout voltage.

In testing, once the oven was warm (a condition in which the OCXO was drawing approximately 250 mA at normal "room temperature") the dropout of the regulator was approximately 50 millivolts - a voltage drop that is a result of the resistance of the wires used to power the unit and the on-resistance of the FET.  This rather simple regulator seems to work quite well, holding the output voltage steady to within a few millivolts over the input voltage range of 11.1 to 17 volts with good transient response.
Figure 4:
The end panel of the OCXO module.  The power feedthrough/capacitor
is on the left, obscured by the red/white power cable with the yellow-ish
"ready" light to the right of it.  The three BNC connectors are the 10 MHz
outputs, allowing multiple devices to be connected while in use and/or while
its calibration is being checked.
Click on the image for a larger version.

"Faster warmup" feature:

This OCXO has a "status" output that, when "cold", outputs about 0 volts and in this state, Q101 is turned off, allowing R112 and R113/D102 to pull its collector high - turning on Q102 - which pulls the gate of Q103 low through R118, turning it fully "on."  In this state the voltage applied to the oven is nearly that of the battery supply and this higher voltage increases the power applied to the oven, allowing it to heat more quickly.  Once the oven's "status" line goes high, Q101 is turned on, illuminating the LED and turning off Q102, allowing the regulator to operate normally.

Note:  When the unit is warming up, the OCXO's voltage is unregulated which means that the supply should be kept below 15.0 volts to stay within the "safe zone" of the ratings of the oscillator itself.

Does the "boosted" voltage actually help the oven warm up faster?  Probably only a little bit, but it took only 4 additional components to add this feature!

Status indicator:

It should be noted that this status line doesn't indicate that the oven has fully warmed up, but only that it's still warming:  At "room temperature" it takes at least another 5 minutes before the frequency will be stable enough for use and another 5 minutes or so after that until it's "pretty close" to the intended frequency and it can be used at microwave frequencies without others having to chase you around.

Why have the indicator light if it doesn't indicate that the unit is actually "ready"?   While this indication isn't perfect if the light isn't on, you can be sure that the frequency output won't be valid for one reason or another.

Because the OCXO itself is somewhat load-sensitive (about +/-1.0x10-9 - perhaps a few 10s of Hz at 24 GHz) U102 - an LM7171 - is used as a distribution amplifier to both isolate the oven from its loads and to provide fan-out to allow multiple outputs to be driven simultaneously.  The LM7171, a high-output, high-speed op amp, is configured for a gain of 2, providing about 2 volts peak-to-peak output with the drive provided by the OCXO.

Mounting the oven:

Because this unit is intended to be used "in the field" it was decided to mount the OCXO module itself to prevent mechanical shock from affecting the reliability, frequency stability and accuracy and this was done using some rubberized mounting pillars from scrapped satellite equipment while some "blobs" of silicone were placed on the wall of the die-cast enclosure to prevent the OCXO housing itself from directly impacting it should the unit be accidentally dropped.

Figure 5:
Schematic of the OCXO-based unit, including the zero-dropout regulator and 10 MHz distribution amplifier.  It is important that the connection of the "ground" side of the 10 turn calibration potentiometer be made at the OCXO and not elsewhere, this to minimize possible frequency shifts due to I*R losses as the oven's heater power changes.
Click on the image for a larger version.
A few bits of stiff foam could also be used to provide some shock mounting in the corners of the OCXO but be aware that some oven-based oscillators have been known to become less accurate and stable if they are over-insulated and can't radiate at least some of their heat, so don't go overboard.


Like any crystal oscillator, it is somewhat "position sensitive" in that a frequency shift of 10s of Hz (at 24 GHz) can be observed if the unit is placed on its side, upside-down, etc. due to the effect of gravity on the quartz crystal itself.  While this effect is very minor, it's worth noting when it's being set to frequency and in operation.

In other words, when you calibrate it (see below) do so in the same physical orientation that it will be when it is in use.

DC input protection and filtering:

The input supply is RF-bypassed using a feedthrough capacitor to prevent the ingress or egress of extraneous RF along the power lead.   For power-supply short-circuit and reverse-polarity protection, R101, a 1.1 amp, self-resetting PTC fuse is used in conjunction with D101, a 3-amp diode.

Why not use a forward-biased diode for reverse-polarity protection?  If you recall, we are going through the trouble of minimizing voltage drop-out with our "special" voltage regulator and we could diminish this if we inserted something that caused a voltage drop - even the 0.3-ish volts of a Shottky diode would undermine this effort.

By using the "reverse-biased diode" and the self-resetting PTC fuse we get:
  • A means of current limiting should something to wrong:  If we accidentally short something out, the fuse resets itself when the fault is cleared - and no need to worry about not having a spare fuse when one is out in the hinterland trying to operate!
  • If the polarity is somehow connected backwards, the diode will conduct and the PTC fuse will "open" - no harm done, returning to normal once the fault is rectified.
  • There is minimal voltage drop related to the fuse as its resistance is a fraction of an Ohm under normal conditions which means that we won't compromise the voltage "headroom" of a 12-volt lead-acid battery.

The best way to calibrate this device is to use a GPS disciplined oscillator or a known-good rubidium frequency reference.  If you have access to one of these, connect the output of the OCXO to one channel of a dual-trace oscilloscope and the known-good frequency reference to the other, triggering on one of two signals - it really doesn't matter which one.

Note:  If you have an analog dual-trace oscilloscope with sufficient bandwidth you can use the "X/Y" mode to produce a Lissajous pattern (obligatory Wikipedia reference here) - but this doesn't always work well on modern, digital scopes when high frequencies are involved due to sample aliasing.

Adjusting the 'scope to see one of the waveforms, one should see a stationary wave (the one on which the 'scope is triggered) while the other will be "sliding" past the first.  Adjust the OCXO's frequency (after the OCXO has warmed up for at least 30 minutes - preferably more) while it is sitting in the same physical orientation in which it will be used as this can (slightly) affect frequency.  To assure a more consistent thermal environment it is suggested that the cover of the enclosure containing this circuitry be left on except during the brief periods to access the 10-turn potentiometer unless provisions are made to access it (via a hole) from outside the box.

The OCXO's frequency is then adjusted to minimize the rate at which the two waveforms are moving with respect to each other:  It's sometimes easier to make this adjustment if the 'scope is adjusted so that the two waves are atop each other and about the same size.  With careful adjustment it should be possible to set the frequency so that the two waveforms that take more than 10 seconds to "slide" past each other - maybe longer.  The Isotemp OCXO should, in theory, be able to hold to that "10 second" slide rate over a wide variety of temperature conditions.

If you don't happen to have access to a rubidium reference or a GPS Disciplined oscillator, you can do "reasonably" well by zero-beating the 10 MHz output with the signal from WWV or WWVH, be note that Doppler shifts can cause their apparent frequencies to shift by 1 Hz or more.  I'll leave the explanation of methods of successfully zero-beating an off-air signal to others on the GoogleWeb.

The best time to attempt this is when you are hearing only one of these two stations (assuming that you can ever hear them both) and when it's signal is the most "solid" - that is, it's fading in and out is at minimum.  Often, the worst time to make this sort of measurement is when any part of the radio path between you and WWV (or WWVH) is within a hour or two of sunrise or sunset as this is when the ionospheric layers are in a state of flux.  If you are hearing both WWV and WWVH, don't try this as the two frequencies and signal strength will not likely be consistent and the results will probably be confusing.

If you don't happen to live in an area where you have a reasonable signal from WWV or WWVH then I suggest you ask around to find someone who has appropriate gear to help with this task.

Comments about alternative schemes for low-dropout regulation for the OCXO:

There are a number of "low-dropout" adjustable regulator ICs on the market that may be suitable for your this project - but there are a few caveats.

For example, there is the Linear Technologies LT1086-Adj which is rated for up to 1.5 amps of current.  While lower dropout than a conventional adjustable regulator such as an LM317, it does have approximately 1 volt of dropout which means that if you set the OCXO's supply voltage to 11.0 volts - the minimum recommended in the OCXO's specification - your battery voltage must be at least 12.0 volts:  While this represents a lead-acid battery that mostly depleted it is likely that a small, but healthy, lead acid could drop to such a voltage under transmit load - particularly if the resistance of power leads is taken into account.  This 3-terminal regulator is used in a manner very similar to the LM317 - except that you really must have some good quality, low-ESR capacitors (probably tantalum) very close to the regulator itself - see the data sheet.

Also made by Linear Technologies is the LT1528 that is rated for up to 3 amps that has a (nominal) 0.6 volts of dropout - more typically in the 0.3 to 0.5 volt area for the amount of current consumed by the OCXO, particularly once it has warmed up:  This extra margin would keep one in the "safe" region of the OCXO's operating voltage range down to around 11.5 volts from the batter allowing both "deeper" discharge and more voltage drop on connecting wires.  This part is somewhat more complicated to use than the LT1086, above, but it is, overall, simpler than the op-amp based regulator described earlier in this page.

If the "fast warmup" were to be implemented on either of the above regulators it would take a different form than the above - likely using several resistors and a transistor or two to "switch" the resistor-programmed voltage setting to something higher than the normal voltage.

There are a number of other, similar, low-dropout regulators that are made by different manufacturers, but very few have as low a dropout voltage (e.g. about 50 millivolts) as the simple FET/Op-amp circuit described on this page.

Additional comments:
  • It is recommended that one not use a switching regulator to power the OCXO unless it has been extremely well filtered and bypassed.  Unless such a regulator is a buck-boost type it will probably have a higher drop-out voltage than even a standard low-dropout linear regulator.  Because of the rather low overhead voltage involved, there is not much loss in the linear regulator - only 10-15% or so with a 12.5 volt supply with a 11.0 volt output - a loss comparable to a garden-variety switching regulator.
  • If you are interested in an example of this project being built with an etched PC board with surface-mount parts, visit VK4ABC's 10 MHz OCXO Web Page.

* * *

This is a revised version of one of my web pages, the original being found at http://www.ka7oei.com/10gig/10meg_oven_1.html


This page stolen from ka7oei.blogspot.com

 Note:  This post is partially an attempt to test means of reducing the "scraping" of content of this blog by sites such as "rssing", who seem to "swipe" content and "load" search engines' result with unwary readers NOT ending up at my page.     xe2XV6SJ9914C50H08S8  QY2IU7TU0C11c57804Q8

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